The present application relates generally to bipolar output stages, and more specifically to a bipolar rail-to-rail class-AB output stage that provides improved AC performance in low voltage applications.
Bipolar class-AB output stages are known that are capable of providing a rail-to-rail output swing. For example, FIG. 1 depicts a conventional bipolar rail-to-rail class-AB output stage 100, which includes transistors Q1–Q4, current sources I1–I2, and a reference voltage generator REF 102 including diode-connected transistors Q5–Q8 and current sources I3–I4. As shown in FIG. 1, the inputs IN1–IN2 of the output stage 100 are applied to the bases of the transistors Q1–Q2, respectively, and the output OUT is provided at the collectors of the transistors Q1–Q2. The NPN-type transistor Q1 and the PNP-type transistor Q2 are configured to form a complementary common emitter stage, and the transistors Q3–Q4 are configured to form a control circuit to assure class-AB operation of the output stage. Reference voltages REF1–REF2 are provided to the bases of the transistors Q3–Q4, respectively, the current source 11 is coupled between the collector of the transistor Q4 and the negative power supply voltage Vee, and the current source 12 is coupled between the emitter of the transistor Q4 and the positive power supply voltage Vcc, thereby biasing of the AB-control circuit formed by the transistors Q3–Q4.
One drawback of the above-described bipolar rail-to-rail class-AB output stage 100 is that the NPN and PNP transistors Q1–Q2 forming the complementary common emitter stage typically have a low current gain β. One way of increasing the overall current gain of the output stage 100 is to employ respective Darlington circuits in place of the single NPN and PNP transistors Q1–Q2. For example, such Darlington circuits may provide current gains equal to about β2 while providing good AC performance. However, relatively large power supplies are generally required for Darlington circuit configurations, thereby making them unsuitable for low voltage applications. Further, because the current gain β of NPN transistors is generally not equal to the current gain β of PNP transistors, the respective current gains β2 of Darlington circuits disposed on the NPN and PNP-sides of the output stage 100 may be significantly different. As a result, positive and negative half-waves provided at the output OUT of the bipolar output stage 100 may be highly unsymmetrical.
FIG. 2 depicts another conventional bipolar rail-to-rail class-AB output stage 200 comprising two cascaded class-AB output stages. The output stage 200 includes the transistors Q1–Q4, the current sources I1–I2, and the reference voltage source REF 102 including the diode-connected transistors Q5–Q8 and the current sources I3–I4. As in the output stage 100 of FIG. 1, the NPN-type transistor Q1 and the PNP-type transistor Q2 are configured to form a first complementary common emitter stage, and the transistors Q3–Q4 are configured to form a first AB-control circuit. The output stage 200 further includes an NPN-type transistor Q1 and a PNP-type transistor Q12 configured to form a second complementary common emitter stage, and transistors Q13–Q14 configured to form a second AB-control circuit. As shown in FIG. 2, the inputs IN1–IN2 of the output stage 200 are applied to the bases of the transistors Q11–Q12, respectively, and the output OUT is provided at the collectors of the transistors Q1–Q2. The reference voltage REF1 is provided to the base of the transistor Q3, the reference voltage REF2 is provided to the base of the transistor Q4, the current source 11 is coupled between the collector of the transistor Q4 and the negative supply voltage Vee, and the current source 12 is coupled between the emitter of the transistor Q4 and the positive supply voltage Vcc, thereby biasing the first AB-control circuit formed by the transistors Q3–Q4. Further, the reference voltage REF1 is provided to the base of the transistor Q13, the reference voltage REF2 is provided to the base of the transistor Q14, a current source 15 is coupled between the collector of the transistor Q14 and the negative supply voltage Vee, and a current source 16 is coupled between the emitter of the transistor Q14 and the positive supply voltage Vcc, thereby biasing the second AB-control circuit formed by the transistors Q13–Q14. The output stage 200 further includes a first current mirror including a diode-connected input transistor Q17 and an output transistor Q16, a second current mirror including a diode-connected input transistor Q19 and an output transistor Q18, capacitors C1–C2 coupled between the output OUT and the input of the first class-AB output stage, and capacitors C3–C4 coupled between the output OUT and the input of the second class-AB output stage, thereby providing frequency compensation by a nested Miller compensation technique.
Although the bipolar rail-to-rail class-AB output stage 200 of FIG. 2 is more suited to low voltage applications than the bipolar output stage 100 of FIG. 1, the output stage 200 also has drawbacks in that the nested Miller compensation technique employed therewith significantly reduces the speed of the circuit. For example, the feedback loop formed by the Miller capacitors C1–C2 and the feedback loop formed by the Miller capacitors C3–C4 each reduce the overall unity gain frequency of the circuit by a factor of about ⅓.
It would therefore be desirable to have an improved bipolar rail-to-rail class-AB output stage that avoids the drawbacks of the above-described conventional bipolar output stages.